Current parking response to transient load demands

ABSTRACT

Embodiments are disclosed relating to an electric power conversion device and methods for controlling the operation thereof. One disclosed embodiment provides an electric power conversion device comprising a first current control mechanism coupled to an electric power source and an upstream end of an inductor, where the first current control mechanism is operable to control inductor current. The electric power conversion device further comprises a second current control mechanism coupled between the downstream end of the inductor and a load, where the second current control mechanism is operable to control how much of the inductor current is delivered to the load.

BACKGROUND

Typical electronic devices, such as microprocessors and the like, mayrequire large changes in input current during operation. As componentscome “online,” current demands may increase dramatically, and similarly,current demands may decrease dramatically as components go “offline.”For example, a Graphics Processing Unit (“GPU”) may utilize a smallamount of current most of the time, but may also require a substantialincrease in current as a new frame is generated and the appropriatecomponents are brought into operation. If such demands occur andsufficient power is not available, the voltage provided to thecomponents may drop below a critical voltage, thus potentially effectingundesirable operation. As such, typical electric power conversiondevices (e.g., voltage regulators) may utilize one or more energystorage devices, such as capacitors and inductors, in order to ensurethat enough energy is available to provide the desired current. However,as the storage devices increase in size, the ability to respond quicklyis proportionally diminished.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 schematically illustrates a typical electric power conversiondevice.

FIG. 2 schematically illustrates an example of an electric powerconversion device according to an embodiment of the present disclosure.

FIG. 3 schematically illustrates an example electronic device comprisingan electric power conversion device according to an embodiment of thepresent disclosure.

FIG. 4 illustrates a process flow depicting an embodiment of a methodfor operating an electric power conversion device.

FIGS. 5 and 6 illustrate example time-domain responses of the exampleelectric power conversion device of FIG. 2.

DETAILED DESCRIPTION

Typical electronic devices, such as computing devices, may be configuredto receive electric power from one or more electric power sources. Forexample, mobile computing devices may be configured to receive electricpower from a battery and/or from a mains power system (i.e., “linepower”), while other devices may be configured to receive electric powerfrom any one or more other sources (e.g., generator, solar panel, etc.).Regardless of the source of the electric power, the electric power istypically delivered at a particular voltage and frequency, and istypically either delivered as alternating current “AC” or direct current“DC.” However, one or more components of a given electronic device maynot be configured to directly utilize the provided electric power. Forexample, although mains electricity in the United States is typicallydelivered as 120V of alternating current at 60 Hz, a particularelectronic device may be configured to utilize 12V DC power. Therefore,the power is converted (e.g., voltage rectified and stepped-down) byvarious mechanisms, or combination of mechanisms, before being utilizedby the electronic device. Accordingly, various approaches utilizing avariety of electric and electronic components exist in order to providesuch power conversion.

However, in order to provide a desired output (e.g., desired frequency,voltage, current, phase, etc.), typical electric power conversiondevices are configured to operate over a range of operating conditions.For example, electricity demanded by a high-performance electronicdevice, such as a CPU or GPU, may fluctuate with time. Specifically, asvarious components of such devices come online (e.g., upon rendering anew frame in a GPU, logic block restarting after a stall, start of alarge computation, etc.), there may be a rapid rise in current demand.Similarly, when one or more components go offline (e.g., upon entering alow-power mode), there may be a rapid fall in current demand. As such,typical electric power conversion devices are configured such that thevoltage level during either of these instances is within a desiredrange.

Turning now to FIG. 1, a typical power conversion device 100 is shown.Device 100 is configured to provide a desired output (e.g., 1V DC) toload 102 (e.g., logic “blocks,” etc.) and capacitor 103 by convertingpower received from electric power source 104 (e.g., battery, mainspower, etc.). For example, by modulating the duty factor of controlsignals 106 and 108 (e.g., PWM signals, PFM signals, etc.), controller110 is configured to selectively enable transistors 112 and 114,respectively. In doing so, controller 110 is able to modulate theaverage current flowing through inductor 116. Specifically, by enablingtransistor 112, the instantaneous current flowing through inductor 116is increased, whereas the instantaneous current is decreased by enablingtransistor 114. The difference between the current flowing through theinductor and the load current is accumulated on capacitor 103. Thus, theoutput voltage provided to load 102 can be controlled by controlling thecurrent through inductor 116.

However, the inductor also resists changes in current, therebypreventing the stored energy in inductor 116 from being released all atonce (e.g., to load 102) when load current changes. This property ofinductors, along with the storage capacity of capacitor 103, enables anoutput voltage at load 102 that is sufficiently stable duringsteady-state operation. Nonetheless, there is some “ripple” in thevoltage at load 102 that depends on the size of inductor 116, the sizeof capacitor 103, and/or the switching frequency of the controller 110,among other factors. Generally speaking, as the size of inductor 116increases, the output ripple at steady state proportionally decreases.Accordingly, inductor 116 may be sized large enough in order to providean output voltage that does not fluctuate outside a desired voltagerange. However, it will be appreciated that the tendency of inductor 116to resist a change in current may be undesirable during a rapid increaseor decrease in current (referred to as “transients”) demanded by load102.

An example configuration of device 100 (e.g., typical 30 A regulatorphase), is as follows. Inductor 116 is 0.5 μH, electric power source 104provides 12V DC, and the desired output to load 102 is 1V DC. Ignoringthe voltage drop across transistor 112 (e.g., due to a small channelresistance) and other non-idealities, the voltage drop across inductor116 is 11V. As such, the maximum (ideal) current response from inductor116, defined as the voltage divided by the inductance, is 22 A/μs.Accordingly, providing an extra 10 A of current to load 102 will take atleast 500 ns, even ignoring other non-idealities (e.g., time tosynchronize control signals 106 and 108 to new demands). While thecurrent being provided is less than the current demanded by load 102,the voltage seen at load 102 will begin to drop as capacitor 103 isdischarged by the current difference. If the voltage drops too far, load102 may operate incorrectly. It will be thus appreciated that suchperformance may be unsatisfactory in some high-performance electronicdevices.

If desired voltage characteristics cannot be satisfied, load 102 may beconfigured to employ various techniques to deal with the providedvoltage. For example, load 102 (e.g., computing device), may beconfigured to “throttle” performance upon detecting a voltage that isoutside, or near an extreme of, the desired voltage range. Throttlingmay include, for example, halting pending operations, decreasing clockfrequency to allow greater time for edge transitions, and/or otherwisedecreasing throughput.

The configuration shown in FIG. 1 is typically referred to as a “buck”converter. While the present invention is described in the context ofthis buck converter, one of ordinary skill in the art will understandthat this invention can be applied to other “switch-mode” powerconversion circuits including, but not limited to, a forward converter,a half-bridge converter, a full-bridge converter, a flyback converter,and/or variants thereof.

FIG. 2 shows an example of an electric power conversion device 200according to an embodiment of the present disclosure. Device 200 isconfigured to provide a desired output at node 202 (e.g., at load 204and capacitor 206) by converting power received from electric powersource 208 (e.g., battery, mains power, etc.). Device 200 comprisesfirst current control mechanism 210 coupled to electric power source 208and an upstream end of inductor 212 (L1). Mechanism 210 is operable tocontrol the average of inductor current 213 (I_(L1)) flowing throughinductor 212. For example, as illustrated, mechanism 210 may include oneor more first switching mechanisms 214 and one or more second switchingmechanisms 216. Mechanisms 214 and 216 may each include, for example,n-type power MOSFETs, and/or other switching mechanisms. Although asingle mechanism 214 and 216 is illustrated for the ease ofunderstanding, it will be appreciated that a plurality of mechanisms 214and/or 216 may be connected in parallel to increase current capacity,decrease conduction losses, etc.

Device 200 further includes controller 218 configured to apply one ormore control signals to the first current control device. For example,controller 218 may be configured to provide first control signal 220 andsecond control signal 222 to mechanism 214 and mechanism 216,respectively. For example, control signals 220 and 222 may include PWM(“Pulse Width Modulation”) or PFM (“Pulse Frequency Modulation) signals,a combination of PWM and PFM, and/or different control signals, in orderto selectively enable mechanisms 214 and 216. Regardless of the specificconfiguration, controller 218 is configured to provide control signals220 and 222 such that both mechanisms 214 and 216 are not concurrentlyenabled. Such a scenario would result in a substantially zero-resistancepath between the supply of electric power source 208 and ground, therebypotentially damaging device 200 and/or resulting in unsuitably highpower usage.

In contrast to the device 100 of FIG. 1, device 200 further comprisessecond current control mechanism 224 coupled between the downstream endof inductor 212 and load 204. In contrast to the first current controlmechanism 210, the second current control mechanism 224 is operable tocontrol how much of inductor current 213 flowing out of the downstreamend of inductor 212 is delivered via supply current 226 (I_(SUPPLY)) tocapacitor 206, and hence to load 204. As such, second current controlmechanism 224 includes one or more third switching mechanisms 228 (e.g.,one or more parallel-connected planar MOSFETs) and one or more fourthswitching mechanisms 230 (e.g., one or more parallel-connected planarMOSFETs). In contrast to mechanisms 214 and 216, the voltage acrossmechanisms 228 and 230 may be substantially less. For example, thevoltage supplied at the downstream of inductor 212 may be substantiallyequivalent to the output voltage at the load. As mechanisms 228 and 230are therefore switching a lesser voltage, mechanisms 228 and 230 may beconstructed from lower-voltage devices, such as “planar” MOStransistors, as compared to mechanisms 214 and 216. Such mechanisms maybe faster than mechanisms 214 and 216, thereby decreasing switchinglosses, and may also be incorporated onto an integrated circuit, therebypotentially reducing space used and/or reducing cost due to the lack ofdiscrete components. For example, mechanisms 228 and 230 may be realizedon the same integrated circuit as load 204, may be integrated on aseparate die or dice on the same package as load 204, or may beintegrated on a separate package. The mechanisms may be realized asstandard-voltage “core” transistors in a typical digitalintegrated-circuit process, or the mechanisms may be realized ashigher-voltage thick-oxide input-output transistors in a typical digitalintegrated-circuit process. In a preferred embodiment, switchingmechanism 230 is a p-type planar MOSFET and switching mechanism 228 isan n-type planar MOSFET. However, one of ordinary skill in the art willunderstand that either type of MOSFET may be used for either mechanismwith appropriate gate-drive circuitry without departing from the scopeof the present disclosure.

Controller 218 may be further configured to apply one or more controlsignals to the second current control mechanism. For example, thecontroller may be configured to provide third control signal 232 andfourth control signal 234 to third switching mechanisms 228 and fourthswitching mechanisms 230, respectively. As with signals 220 and 222,signals 232 and 234 may utilize PWM, PFM, and/or any other suitablecontrol schema in order to selectively enable mechanisms 228 ormechanisms 230. In some embodiments, signals 232 and 234 may be at leastpartially synchronous with signals 220 and 222. In other embodiments,the signals may be provided asynchronously. Furthermore, signals 228 and230 may be provided (e.g., modulated) at a different frequency thansignals 220 and 222 in some embodiments.

Regardless of the specific configuration of the control signals,controller 218 may be configured to selectively enable third switchingmechanisms 228 and disable fourth switching mechanism 230 to disablesupply current 226. Specifically, by enabling mechanisms 228, all of theinstantaneous inductor current 213 flowing through inductor 212 isdiverted through mechanisms 228 to ground instead of being delivered tocapacitor 206. Conversely, by enabling fourth switching mechanisms 230and disabling third switching mechanism 228, substantially all of theinstantaneous inductor current 213 flowing through inductor 212 (minustransistor conduction losses, inductor winding resistance, etc.) isprovided to capacitor 206. The duty factor (D) of mechanism 230determines the fraction of inductor current (I_(L)) in inductor 212 thaton average is supplied to capacitor 206. In turn, capacitor 206 smoothesthe square wave supply current 226 waveform to generate load current 235(I_(LOAD)) according to the duty factor and the inductor current, asfollows: I_(LOAD)=D×I_(L). As with mechanisms 214 and 216, controlsignals 232 and 234 are provided such that both mechanisms 228 and 230are not concurrently enabled to avoid a short circuit across capacitor206.

Until a need for a current transient is anticipated (e.g., duringsteady-state operation), device 200 operates in a manner substantiallysimilar to device 100. That is, mechanism 228 is disabled and mechanism230 is enabled such that substantially all of inductor current 213 isprovided to node 220 as supply current 226. Further, mechanisms 214 and216 (e.g., “switches”) are selectively enabled (“switched”) in order tocontrol current 213, and hence the voltage seen by load 204. In thisway, if the load voltage is constant, supply current 226 issubstantially equivalent to the inductor current.

Briefly turning now to FIG. 5, an example time-domain response 500 isshown for device 200 in steady-state operation. As illustrated, FIG. 5includes waveforms 502 and 504 illustrating the response of mechanisms220 (“M1”) and 222 (“M2”), respectively (e.g., response effected bytime-varying control signals 220 and 222). Specifically, waveforms 502and 504 alternate between complementary “ON” and “OFF” states. Waveform506 in turn illustrates the voltage (“V1”) at the upstream end ofinductor 212 that is controlled by the switching of mechanisms 220 and222. As described above and as illustrated by waveform 506, voltage V1is high (nominally equal to V_(SUPPLY)) when first switching mechanism214 is in an “ON” state, and is low (nominally 0V) when second switchingmechanism 216 is an “ON” state. For example, when mechanism 214 is “ON”(i.e., conducting), mechanism 214 ideally provides a zero-resistancepath between the output from electric power source 208 and the upstreamend of inductor 212. Similarly, when mechanism 216 is “ON,” mechanism216 ideally provides a zero-resistance path between the upstream end ofinductor 212 and ground. Therefore, waveform 506 toggles between a firststate where voltage V1 is substantially equivalent to the input voltageand a second state where voltage V1 is substantially zero (i.e., atground).

Continuing with response 500, waveform 508 illustrates the time-domainresponse of inductor current 213 in inductor 212. As inductors resistchanges in current, waveform 508 slowly “ramps up” when mechanism 214 isenabled, and slowly “ramps down” when mechanism 216, thereby potentiallyproducing the illustrated sawtooth waveform 508. Waveform 508 is furtherillustrated with a representation 510 of the average inductor current213 provided by the sawtooth waveform.

Finally, FIG. 5 further includes waveform 512 illustrating the responseof the voltage (“VC1”) at output node 202 (e.g., voltage acrosscapacitor 206 and seen by load 204). Specifically, as current 213increases in inductor 212, charge accumulates on capacitor 206, therebyinducing a voltage across the capacitor. As current 213 ramps down,capacitor 206 supplies the accumulated charge to power the load, andthus the voltage VC1 decreases as the charge is depleted. As capacitor206 oscillates between these “charged” and “depleted” states, voltageripple 514 about average voltage 516 is produced. As previouslymentioned, the size of capacitor 206 may be increased, and/or otheradjustments may be made, in order to decrease ripple 514.

In the above-described operating mode, it will be appreciated that theability of device 200 to respond to transients is similar to that ofdevice 100. Namely, the size of the inductor, and thus the property ofresisting changes in current, determines how quickly device 200 canrespond to a transient demand.

Returning to FIG. 2, in order to provide improved transient handling,for example, controller 218 and current control mechanism 224 may beconfigured to control how much of inductor current 213 flowing throughinductor 212 is provided to satisfy the operating requirements of load204. This may be achieved by applying control signals to current controlmechanism 224 to place the mechanism into any of a plurality of states.Each such state may be characterized by how much of the availablecurrent (i.e., inductor current 213) is provided to the load (e.g., 50%,80%, etc.). In the depicted configuration, the states may also becharacterized by a duty factor used to control one or more switchingmechanisms of current control mechanism 224. In other words, a givenduty factor may be used to provide a certain percentage of availablecurrent, with the duty factor being increased or decreased torespectively increase or decrease the current percentage supplied to theload. The term “states” does not necessarily imply discrete states; inthe depicted configuration, the control signals may be controlled toprovide any desired percentage of inductor current 213 to the load. Thismethod of operation, in which inductor 212 is charged to provide anavailable current which is then passed on to the load in a controllablepercentage, may be referred to as a “current parking” configuration, ora configuration in which current is “reserved” via inductor 212.

Starting with first current control mechanism 210, current parking mayproceed as follows. Current control mechanism 210 is controlled bycontroller 218 to control the charging of inductor 212 so as to “park”or “reserve” a desired amount of inductor current 213. In the depictedexample, this is achieved by controlling the states of switchingmechanisms 214 and 216 with signals 220 and 222. The state of theswitching mechanisms is rapidly toggled between two states: (1) switch214 is closed and switch 216 is open—thereby increasing the current inthe inductor; and (2) switch 214 is open and switch 216 isclosed—thereby decreasing the current in the inductor by applying avoltage opposing the flow of current. Duty factor may be controlled viaPWM, PFM or other appropriate schemes to achieve the desired conditionsat the downstream end of the inductor. Typically, the control will beperformed so that the average of inductor current 213 is greater thanthe average current needed by the load at steady state operation. Thisexcess energy is stored in inductor 212 in order to rapidly andefficiently respond to transient current demands.

Second current control mechanism 224 is then controlled by signals fromcontroller 218 to provide a desired amount of the parked current to theload. Similar to first current control mechanism 210, the second currentcontrol mechanism is rapidly toggled between a state in which inductorcurrent 213 is delivered to the load, and a state in which the inductorcurrent is discharged along a path that bypasses the load (e.g., a pathto ground). This toggling is achieved in the depicted example bycontrolling third switching mechanism 228 and fourth switching mechanism230 with control signals 232 and 234, respectively. Specifically, loadvoltage across capacitor 206 is increased by closing switch 230 andopening switch 228; load voltage across capacitor 206 is decreased byopening switch 230 and closing switch 228. By controlling the dutyfactor of the switches (e.g., via PWM, PFM or other schemes), a desiredaverage percentage of the parked current is provided to the load. Asmentioned previously, this duty factor is set so that the average supplycurrent 226 supplied to capacitor 206 is identical to the load current,such that the load current is equal to the duty factor multiplied by theload current and the voltage at node 202 (e.g., voltage across capacitor206 and hence the load voltage) is held constant.

Briefly turning now to FIG. 6, an example time-domain response 600 isshown for device 200 providing “current parking.” As illustrated, FIG. 6includes waveforms 602 and 604 illustrating the response of mechanisms228 (“M3”) and 230 (“M4”), respectively (e.g., response effected bytime-varying control signals 232 and 234). Specifically, waveforms 602and 604 alternate between complementary “ON” and “OFF” with a particularduty factor in order to set the voltage (“V2”) at the downstream end ofinductor 212.

Waveform 606 in turn illustrates voltage V2, which is high (nominallyequal to VC1) when first switching mechanism 214 is in an “ON” state,and low (nominally zero) when fourth switching mechanism 230 is an “ON”state. For example, when mechanism 230 is “ON” (i.e., conducting),mechanism 230 ideally provides a zero-resistance path between theupstream end of inductor 212 and node 202. Similarly, when mechanism 228is “ON,” mechanism 228 ideally provides a zero-resistance path betweenthe upstream end of inductor 212 and ground. Therefore, waveform 606toggles between a first state where voltage V2 is substantiallyequivalent to the output voltage (e.g., voltage “VC1” across capacitor206) and node 202 and a second state where voltage V2 is substantiallyzero (i.e., at ground).

As illustrated by waveform 608, inductor current 213 may be maintainedat a constant level (e.g., by controlling first current controlmechanism 214). However, by switching second current control mechanism224 according to the schema illustrated by waveforms 602 and 604, a loadcurrent may be provided that is less than the inductor current.Accordingly, waveform 610 illustrates the response of supply current226. When fourth switching mechanism 230 is conducting, supply current226 may be substantially equivalent to inductor current 213 (e.g., lessleakage currents, etc.). Similarly, when third switching mechanism 228is conducting, supply current may be substantially zero. Load current235 (I_(LOAD)), as illustrated at 612, is an intermediate value betweenthe extremes of supply current 226. In the periodic steady state, theload current is equivalent to the average of the supply current.

Finally, response 600 includes waveform 614 illustrating the response ofthe voltage (“VC1”) at output node 202 (e.g., voltage across capacitor206 and seen by load 204). Specifically, when mechanism 230 is ON andinductor current is larger than load current, charge accumulates oncapacitor 206, thereby increasing the voltage across the capacitor. Whenmechanism 228 is ON, capacitor 206 supplies the accumulated charge topower the load without receiving any input current, and thus the voltageVC1 decreases as the charge is depleted. As capacitor 206 oscillatesbetween these “charged” and “depleted” states, a saw-tooth voltageripple 616 about average voltage 618 is produced. Due to the toggling offourth switching mechanism 230, waveform 614 comprises a sawtooth shapein contrast to the sinusoidal ripple 514 illustrated in FIG. 5.

From the above discussion, it will be appreciated that by producing anexcess current and selectively controlling its delivery, the depictedconfiguration can rapidly respond to transient current demands from theload. Without the depicted configuration, response time would be muchslower due to the time required to vary the current through theinductor. Specifically, the ability of device 200 to respond totransients is substantially determined by the switching speed ofmechanisms 228 and 230. The presence of this very fast control loopalleviates concerns about the slow control loop used to control theinductor current. Accordingly, inductor 212 can be sized in order toprovide the desired amount of parked current without concern for theability of the inductor to respond rapidly to current transients.Varying the duty factor at current control mechanism 224 is more or lessinstantaneous, and in any case is dramatically faster than creating avariation in the inductor current. Further, as mentioned above, it willbe appreciated that the lower voltages on the downstream side of theinductor may allow use of transistor types that provide relativelyfaster switching as compared to transistors of the first current controlmechanism. For example, the lower voltages may allow use of planartransistors so as to provide, for example, a control response in theneighborhood of 1 to 10 ns. Such a response may be substantially quickerthan the response of inductor 212 (e.g., 500 ns for a 0.5 μH inductor toprovide an extra 10 A of current). In some embodiments, such a controlresponse may be made faster than the resonant frequency of a typicalintegrated circuit package. Further, the ability to quickly switchmechanisms 228 and 230 may also enable the size of capacitor 206 to bereduced, often by several orders of magnitude. Therefore, in many cases,the capacitor is small enough such that it may be realized entirely onthe integrated-circuit package.

In one embodiment of the present invention, current control mechanism224 is controlled by a periodic control method, such as PWM, PFM, etc.or some combination thereof. In this case, the time to respond to achange in load current may be as large as one period of the periodiccontrol (e.g., 10 ns for a 100 MHz control). In other embodiments, thecurrent control mechanism may be switched near-instantaneously (i.e.,turn mechanism 230 on and turn mechanism 228 off) to supply current tothe load as soon as increased load current (and/or reduced load voltage)is sensed. In other words, in such embodiments, device 200 may be ableto respond to changes in demand as fast as controller 218 is able todetect the need for more current and switch mechanisms 230 and 228. Insuch embodiments, the response may typically take less than ins,depending on the sensing mechanism.

In addition to an increased demand for current by load 204, device 200may be further configured to respond to a decreased demand for currentby load 204 (i.e., a negative current transient). Specifically, whenload 204 stops consuming current or consumes a lesser amount of current,controller 218 can respond substantially instantaneously by reducing theduty factor(s) employed at switching mechanisms 228 and 230.

It will be appreciated that, in other embodiments, device 200 maycomprise a single “phase” of a multi-phase electric power conversiondevice. In such scenarios, additional current may be provided at node202 by one or more other phases. In other words, each individual phasemay be configured to provide a subset of the overall current (e.g., eachphase provides a portion of the current at a different phase), therebypotentially decreasing the size (e.g., due to decreased power handling)of various components of the electric power conversion device.

However, various configurations including multiple phases are possible.For example, in some embodiments, a first phase may provide typicalvoltage conversion (e.g., via device 100 of FIG. 1) during“conventional” device operation. The second phase may have aconfiguration similar to that of device 200, and the second phase maynormally operate by parking all the current in the correspondinginductor without delivering any current to the load. Upon recognizing anincreased current demand, the second phase may be operable to providethe parked current to the load until the transient subsides, and maythen return to the previous operating mode. It will be appreciated thatthese scenarios are presented for the purpose of example and that one ormore stages of a multi-phase electric power conversion device mayutilize current parking according to additional and/or different schemasand via various circuit topologies without departing from the scope ofthe present disclosure.

In order to provide the above-described current delivery and response,controller 218 may be configured to monitor various components and/ornodes of device 200. For example, controller 218 may be configured toreceive node voltage 236 (e.g., voltage input from electric power source208) and/or node voltage 238 (e.g., output voltage at node 202).Controller 218 may incorporate such information into one or more controlloops and/or different logic in order to effect modulation of controlsignals 220, 222, 228, and 230. In some embodiments, controller 218 maybe configured to estimate one or more currents (e.g., inductor current213, load current 235, etc.) from the measured voltage(s). In otherembodiments, controller 218 may be configured to measure one or morecurrents via various current sensing mechanisms. It will be appreciatedthat these scenarios are presented for the purpose of example, andcontroller 218 may be configured to measure and/or estimate any one ormore voltages and/or currents within device 200 in order to provide suchmodulation without departing from the scope of the present disclosure.

However, it will be appreciated that by simply monitoring othercomponents of device 200, controller 218 may not detect future currentdemands sufficiently ahead-of-time in order to park current in inductor212. Thus, in some embodiments, controller 218 may be configured toprovide “parked” current all the time in order to ensure that thedemanded current is always available. However, as current parkingeffects some power losses (e.g., due to switching losses, “on”resistance across transistor channels, etc.), such a configuration maynot provide suitable performance in some scenarios (e.g., mobile devicescenarios or other low-power applications).

As such, in other embodiments, controller 218 may utilize one or morelearning mechanisms in order to anticipate future demands. For example,controller 218 may learn that the current demanded from load 204 dropssubstantially below an average observed load current immediatelypreceding a large increase in demand. For example, if load 204 is a GPU,the load current demand may drop substantially in an idle period beforea new frame is rendered. Accordingly, upon detecting such a drop-off incurrent, controller 218 may be configured to park current in inductor212 according to previously-observed current demands. Thus, whenrendering occurs and the appropriate components are brought online,suitable current is available. It will be appreciated that suchscenarios are presented for the purpose of example, and are not intendedto be limiting in any manner.

However, in some embodiments, it may be desirable to provide moreintelligent control over the operation of device 200. For example,turning now to FIG. 3, an electronic device 300 (e.g., computing device)comprising an electric power conversion device 302 according to anembodiment of the present disclosure (e.g., electric power conversiondevice 200) is illustrated. In some embodiments, device 300 may includeelectric power source 304 (e.g., internal battery) and/or may beoperatively coupled to one or more external electric power sources 306(e.g., mains power).

Device 300 comprises a plurality of components 308, illustrated as anarbitrary number N of components (e.g., logic blocks, discretecomponents, etc.). Some components of device 300, for example “Component1,” may be configured to interface directly with one or more of electricpower source 304 and 306. Component 1, for example, may be configured tooperate from the voltage supplied by the power source(s) and/or mayinclude one or more internal regulation mechanisms. Other components ofdevice 300, for example “Component 2” may be operatively coupled topower conversion device 302. In addition to receiving electric powerfrom device 302, such components may be configured to interact withcontroller 310 (e.g., controller 218) of electric power conversiondevice 302. For example, controller 310 may be configured to monitor oneor more nodes of the components in order to anticipate future currentdemands in a manner similar to the monitoring the “local” nodesdescribed above in reference to FIG. 2. In other words, a change in nodevoltage or current may provide a “trigger” to controller 310 in order toeffect current parking within electric power conversion device 302.

In some embodiments, the components 308 interacting with controller 310may be configured to provide a trigger (e.g., one or more signals) tocontroller 310 alerting the controller of future current demands. Forexample, upon leaving a low-power mode, one or more components 308 maybe configured to provide a signal to controller 310 alerting thecontroller that an increase in load current may soon be needed.Accordingly, controller 310 may be configured to start parking currentupon receipt of such trigger. As another non-limiting example, one ormore components 308 (e.g., processing pipeline component(s)) may beconfigured to provide a signal to the controller upon receipt aparticular architectural instruction within a processing pipeline, suchas the fetching of a floating point instruction, that will require idleexecution mechanisms to come online in a few clock cycles.

It will be appreciated that these triggers are presented for the purposeof example, and that the electric power conversion device described byexample hereto may be configured to respond to any suitable trigger, orcombination of triggers, without departing from the scope of the presentdisclosure.

Turning now to FIG. 4, an example method for providing power to a load,such as a piece of computing logic, and for responding to transientcurrent demands of the load is illustrated. As shown at 402, the methodincludes delivering energy from a power source to an inductor so thatthe average current flowing in an inductor is higher than the averagecurrent needed by a load that is operatively connected to the downstreamend of the inductor. At 404, the method further includes using a currentcontrol mechanism to control how much of the inductor current isdelivered to the load. The amount of current passed to the load may benone, some or all of the available inductor current. As described above,the available inductor current may be referred to as “parked” or“reserved” current, which alludes to the characteristic that thiscurrent is in excess of the average current needed by the load, theexcess being available for use in the event of a future transientincrease in the amount of current needed by the load.

As in the above hardware examples, the inductor may be charged using acurrent control mechanism having switches that are duty-cycle-controlled(e.g., via PWM, PFM, etc.) to achieve the desired inductor currentand/or voltage. Further, a current control mechanism, such as mechanism224 (FIG. 2), may be employed to control how much of the parked inductorcurrent is passed along to the load. That said, it will be appreciatedthat inductor charging, controlled transmission of inductor current,and/or any other functionality described in method 400 may be achievedusing hardware other than that described above.

Variation in the amount of delivered current may be precipitated byrecognizing (sensing) a change in the amount of current needed by theload. This may be an implicit sensing resulting from a voltage drop orother sensed phenomena at the node between the inductor and the load.Alternatively, or additionally, another component or process may signala controller to indicate that current demands are about to change. Therecognizing of a change in the amount of current needed by the load isshown in the example method at 406.

Steps 408 and 410 depict an example of responding, respectively, to anincreased demand for load current and a decreased demand for loadcurrent. These examples suppose a switching arrangement in which a dutyfactor is varied to satisfy the need for a different amount of current.At 408, an increased need is provided by increasing the duty factor of aswitch coupled between the downstream end of the inductor and the load,and decreasing the duty factor of a switch in a current discharge pathfrom the inductor that bypasses the load. In other words, the dutyfactors of the switches are varied so that more of the parked current isdelivered to the load, and less of the parked current is dumped toground. Alternatively, at 410, a negative current transient (reducedneed) is serviced by decreasing the duty factor of the switch betweenthe inductor and the load, and by increasing the duty factor thatcouples the inductor current into the discharge path.

Aspects of this disclosure have been described by example and withreference to the illustrated embodiments listed above. Components thatmay be substantially the same in one or more embodiments are identifiedcoordinately and are described with minimal repetition. It will benoted, however, that elements identified coordinately may also differ tosome degree. The claims appended to this description uniquely define thesubject matter claimed herein. The claims are not limited to the examplestructures or numerical ranges set forth below, nor to implementationsthat address the herein-identified problems or disadvantages of thecurrent state of the art.

1. An electric power conversion device, comprising: a first currentcontrol mechanism coupled to an electric power source and an upstreamend of an inductor, where the first current control mechanism isoperable to control inductor current; and a second current controlmechanism coupled between the downstream end of the inductor and a load,where the second current control mechanism is operable to control howmuch of the inductor current is delivered to the load.
 2. The electricpower conversion device of claim 1, further comprising a controllerconfigured to provide one or more control signals to the first currentcontrol mechanism and the second current control mechanism, so as tocontrol inductor current and delivery of some or all of that inductorcurrent to the load.
 3. The electric power conversion device of claim 2,where the first current control mechanism includes one or more switchingmechanisms that are controlled by control signals from the controller toselectively increase and decrease current flowing through the inductor.4. The electric power conversion device of claim 3, where the one ormore switching mechanisms of the first current control mechanism includeone or more parallel-connected power MOSFETs that are controlled by thecontroller using one or more of a pulse width modulation (PWM) signaland a pulse frequency modulation (PFM) signal.
 5. The electric powerconversion device of claim 2, where the second current control mechanismincludes one or more switching mechanisms that are controlled by controlsignals from the controller to selectively increase and decrease currentprovided to the load.
 6. The electric power conversion device of claim5, where the one or more switching mechanisms of the second currentcontrol mechanism include one or more parallel-connected planar MOSFETs.7. The electric power conversion device of claim 5, where the one ormore switching mechanisms of the second current control mechanisminclude a first switch coupled between the downstream end of theinductor and the load, and a second switch coupled between thedownstream end of the inductor and a current path that bypasses theload, and where the controller causes current flowing through theinductor to be selectively (i) delivered from the inductor to the loadby closing the first switch and opening the second switch, and (ii)diverted away from the load by opening the first switch and closing thesecond switch.
 8. The electric power conversion device of claim 7, wherethe first switch and the second switch of the second current controlmechanism are controlled by signals that include one or more of a pulsewidth modulation (PWM) signal and a pulse frequency modulation (PFM)signal.
 9. The electric power conversion device of claim 7, where thecontroller is configured to recognize a need for an increase in currentdelivered to the load and respond to such need by varying controlsignals provided to the first switch and the second switch of the secondcurrent control mechanism.
 10. The electric power conversion device ofclaim 9, where recognizing the need for an increase in current deliveredto the load includes one or more of observing a negative voltagetransient at the load and receiving an external control signal receivedfrom a device operatively coupled to the controller.
 11. The electricpower conversion device of claim 9, where the controller is furtherconfigured to recognize a need for a decrease in current delivered tothe load and respond to such need by varying control signals provided tothe first switch and the second switch of the second current controlmechanism.
 12. The electric power conversion device of claim 2, wherecontrol signals applied to control the first current control mechanismare modulated at a different frequency than control signals applied tocontrol the second current control mechanism.
 13. An electric powerconversion device, comprising: a first current control mechanism coupledto an electric power source and an upstream end of an inductor, wherethe first current control mechanism includes a first switching mechanismand a second switching mechanism operable to selectively control averageinductor current; and a second current control mechanism coupled betweenthe downstream end of the inductor and a load, where the second currentcontrol mechanism includes a third switching mechanism and a fourthswitching mechanism operable to control how much of the inductor currentis delivered to the load.
 14. The electric power conversion device ofclaim 13, further comprising a controller configured to (i) applycontrol signals to the first switching mechanism and the secondswitching mechanism to control the average current flowing through theinductor; and (ii) apply control signals to the third switchingmechanism and the fourth switching mechanism to control how much of theinductor current is delivered to the load.
 15. The electric powerconversion device of claim 14, where the controller is configured tocontrol the switching mechanisms so that (i) the first switchingmechanism and the second switching mechanism are not concurrentlyenabled; and (ii) the third switching mechanism and the fourth switchingmechanism are not concurrently enabled.
 16. The electric powerconversion device of claim 14, where the controller varies duty factorsof the third switching mechanism and the fourth switching mechanism inresponse to a change in current needed by the load.
 17. A method ofcontrolling current delivered to a load, comprising: controlling energydelivered from an electric power source to an inductor, so as togenerate a desired average inductor current; and controlling a currentcontrol mechanism operatively coupled between the downstream end of theinductor and the load, where such controlling is performed to controlhow much of the inductor current is provided to the load.
 18. The methodof claim 17, where the current control mechanism includes a firstswitching mechanism coupled between the downstream end of the inductorand the load, and a second switching mechanism coupled within a currentdischarge path beginning at the downstream end of the inductor and thatbypasses the load.
 19. The method of claim 18, further comprisingresponding to an increase in load current demand by doing one or both ofincreasing a duty factor of the first switching mechanism and decreasinga duty factor of the second switching mechanism.
 20. The method of claim19, where controlling energy delivered from the electric power source tothe inductor includes generating the desired average current to a levelthat is higher than the anticipated average current needed by the load.